Introduction

Fano resonance has received much attention due to the interesting physics such as distinctly asymmetric shape and high quality-factor (Q-factor)1. The interference of a discrete autoionized state with a continuum gives rise to characteristically asymmetric peaks in excitation spectra, which can be extended to the resonance scattering of quantum theory2,3,4. Recently, the classical oscillator systems enabled by plasmonic nanostructures and metamaterials have led to the achievement of asymmetric Fano-type transmission/reflection in the optical frequencies, which has open up a new perspective towards achieving high-precision nanoscale sensors5,6,7,8,9. Furthermore, the steep Q-factor profile promises applications in bio/chemical sensors10,11,12,13.

A discrete autoionized state and a continuum can be analogue of a broadband-bright mode and a narrowband-dark mode depending on the coupled approach with the incident light from free space14. The bright mode has a large scattering cross section and a low quality factor due to the radiation coupling, which is always excited directly by external energy. On the contrary, the dark mode normally has a significantly larger quality factor, which is only limited by the loss performance and excited indirectly by the bright mode15. The formation exhibits the interference phenomena, where constructive interference corresponds to resonant enhancement and destructive interference to resonant suppression of the transmission16,17. Furthermore, The circuit system which is an effective-mapping image of the classical mechanics can be devoted to the mechanism of Fano resonance18. In passive electric system, the inductance represents a behavior increasing with higher frequency in spectra domain and the capacitance effects the opposite process. Accordingly, the electric resonance is the equilibrium state when the functionality of inductance and capacitance are balance. Based on this, the electric-dynamic equations of the ‘bright’ and ‘dark’ electric-resonant modes are established and imitate Fano resonance effectively. Nevertheless, the circuit structures of high-Q-factor resonances consist of numerous orders of electric resonances and the solutions of dynamic-differential equations are extremely complicated. Therefore, the ultimate goal of simple structure and an effectively steady-convenient analysis of Fano-like resonance are highly desired.

In this paper, we formulate the series and parallel circuits consisting of inductors and capacitors for various-resonant modes and the resistor represents the damping of the oscillators. Additionally, we propose the stable-input impedance mechanism of passive circuit system to mimic the functionality of the Fano resonance. Based on this theory, the pole-zero adjustment of the input impedance can implement arbitrary Q-factor asymmetric resonance flexiblely in simple circuit system which consists of only three passive components (such as two inductors and one capacitor). Furthermore, various resonances (such as Lorentz-like and reversely EIT formations) can be imitated by four electric components. This approach has well-defined Fano-like effective properties and opens up the possibilities to construct extremely high-Q-factor devices while maintaining the simplification of the system. Besides, they can be a guidance for design in microwave or optical circuits and, in particular, for periodic artificial electromagnetic materials. Additionally, it is interesting to note that the passive circuit approach and the theoretical propositions presented in this work processes to achieve high-precision and compressed-composition components19.

Fano resonances without damping

First, we consider Fano resonances without damping, where the circuits are schematically shown in Fig. 1(a). We calculate the stable-input impedance of the circuit which is embedded in the single-input-single-output (SISO) system20 and tune its pole-zero21. The transmittance is defined as S21 = Poutput/Pinput, where Pinput and Poutput are the incident and transmitted power, respectively. The stable-input impedance of series inductor-capacitor (LC) circuit and parallel LC circuit are given as

Figure 1
figure 1

(a) The schematic of the series and parallel-LC circuit as the branch parallel in the main-energy thread. Here the inductors are Ls = Lp = 1.0132 nH and the capacitors are Cs = Cp = 1 pF. (b) The transmittance S21 of the series and parallel-LC circuit system, for the series-circuit branch parallel in system, the reflected-resonant frequency and the parallel-circuit branch parallel in system, the transparent-resonant frequency . (c) The transmittance of series circuit with various Ls = 5 nH, 3 nH, 1 nH and the corresponding capacitor Cs = 0.2026 pF, 0.3377 pF, 1.1032 pF containing the resonant frequency ωs0 = 5 GHz. (d) The transmittance of parallel circuit with various Cp = 5 pF, 3 pF, 1 pF and the corresponding inductor Lp= 0.2026 nH, 0.3377 nH, 1.1032 nH containing the resonant frequency ωp0 = 5 GHz.

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and

respectively, where the resonant frequencies depend on the inductor Ls/p = 1.0132 nH and the capactor Cs/p = 1 pF. In Eq. (1), the stable-input impedance of series LC circuit Zin_series has the zeros ω = ±ωs0 and the poles ω = 0. Here, the negative frequency ω = −ωs0 are ignored due to its physical-meaningless. Then, the input impedance of series circuit is shorted to the ground at the zero ω = ωs0, which leads the input energy total reflected and the transmittance is lowest S21 = 0 as the solid line in Fig. 1(b). In Eq. (2), the input-impedance function of parallel LC circuit has the zero ω = 0 and the poles ω = ±ωp0. Excluding the physical-meaningless pole ω = −ωp0, the input-impedance of parallel LC circuit is infinite at the pole point ω = ωp0 and the transmittance is all-pass S21 = 1 as the dashed line in Fig. 1(b). From Eq. (1, 2), the steep in the vicinity of ωs0 and ωp0 is proportional to the inductor Ls in the series circuit and inversely proportional to the capacitor Cp in the LC-parallel circuit. Therefore, the Q factor can be adjusted by the inductor Ls and the capacitor Cp as shown in Fig. 1(c,d), meanwhile, the corresponding capacitor Cs and inductor Lp is modified for the remaining of resonant frequency ω0s/p = 5 GHz.

Here the Q-factor is expressed as Q = ω0/(ωH − ωL), where ω0 is the central resonant frequency and ωL, ωH are the half-amplitude frequencies lower and higher than ω0. In Fig. 1(c), the series-LC Q-factor are 10.8, 6.5 and 2 for the various inductor Ls = 5 nH, 3 nH and 1 nH. which presents the series-LC resonance sharper with decreasing series inductor Ls. In Fig. 1(d), the parallel-LC Q-factor are 2.27, 1.36 and 0.45 for the various capacitor Cp = 5 pF, 3 pF and 1 pF, which presents the parallel-LC resonance sharper with increasing parallel capacitor Cp.

Based on the above analysis, we can build the Fano-like asymmetric resonance by a series-LC circuit which represents the narrowband-dark mode coupling with a capacitor or an inductor as the broadband-bright mode, as shown in Fig. 2(a,d). Here we use the stable-input impedance method instead of oscillators-dynamic equations in spectra domain to reveal the mechanism of the asymmetric-coupling modes. In Fig. 2(a), the complementary capacitor Cc is added parallel to the series-LC resonance and the the stable-input impedance of this circuit system is:

Figure 2
figure 2

(a) The schematic of the complementary capacitor Cc parallel in the series LC circuit. (b) The transmittance of the circuit system in Fig. 2(a) with different complementary capacitor Cc = 20 pF, 50 pF and 100 pF. (c) The transmittance of the circuit system in Fig. 2(a) with different series capacitor Cs = 0.1 pF, 0.5 pF and 1 pF. (d) The schematic of the complementary inductor Lc parallel in the series LC circuit. (e) The transmittance of the circuit system in Fig. 2(d) with different complementary inductor Lc = 0.1 nH, 0.05 nH and 0.01 nH. (f) The transmittance of the circuit system in Fig. 2(d) with different series inductor Ls = 5 nH, 15 nH and 20 nH.

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Abandoning the physical meaningless solutions, we get the pole of stable-input impedance in Eq. (3) which is greater than the zero ωs0. In addition, the zeros and poles are corresponding to the reflect and transparent resonant frequencies respectively in main-energy thread. Therefore, the transmittance can steep down to zero ωs0 at the higher-frequent pole with the coefficient and presents the formation of Fano-like asymmetric resonance and and high-Q factor. Further, we can get the infinite-Q-factor by turning the pole greatly close to the zero through increasing the complementary capacitor Cc and decreasing the series capacitor Cs. Here we maintain the series-resonant frequency ωs0 = 5 GHz and increase the complementary capacitor Cc = 20 pF, 50 pF, 100 pF, that leads to the transparent resonance 5.132 GHz, 5.050 GHz and 5.025 GHz closing to the reflect resonance ωs0 = 5 GHz gradually and the resonance becomes sharper, as shown in Fig. 2(b). When the complementary capacitor Cc = 20 pF and decreasing the series capacitor Cs = 1 pF, 0.5 pF, 0.1 pF, under the conditions of the series inductor Ls changing correspondingly for maintaining the series-resonant frequency ωs0 = 5 GHz, the transparent resonance is 5.132 GHz, 5.062 GHz, 5.013 GHz closing to the reflect resonance ωs0 = 5 GHz gradually and the Q-factor becomes higher, as shown in Fig. 2(c).

The complementary inductor Lc is parallel-added in the series-LC circuit, as shown in Fig. 2(d) and the the stable-input impedance is:

Abandoning the physical meaningless solutions, we get the pole in Eq. (4) lower than the zero ωs0. Therefore, the transmittance can steep down to zero at the pole located lower than the zero ωs0 when the coefficient and presents the formation of Fano-like asymmetric resonance. Further, we can get the infinite-Q-factor by turning the pole point close to the zero point through decreasing the complementary inductor Lc and decreasing the series inductor Ls. Here we maintain the series-resonant frequency ωs0 = 5 GHz and decrease the complementary inductor Lc = 0.1 nH, 0.05 nH, 0.01 nH, that leads the transparent resonance 4.770 GHz, 4.881 GHz, 4.976 GHz closes to the reflect resonance ωs0 = 5 GHz gradually and the Q-factor becomes higher, as shown in Fig. 2(e). When the complementary capacitor Lc = 0.1 nH is constant and increasing the series inductor Ls = 5 nH, 15 nH, 20 nH, under the conditions of the series inductor changing correspondingly for maintaining the series-resonant frequency ωs0 = 5 GHz, the transparent resonance is 4.951 GHz, 4.976 GHz, 4.986 GHz closing to the reflect resonance ωs0 = 5 GHz gradually and the Q-factor becomes higher, as shown in Fig. 2(f).

In Fig. 2(b), the Q-factor is 2513, 1263 and 197 for Cc = 100 pF, 50 pF and 20 pF, which presents the resonance sharper with the increasing the complementary capactor Cc. In Fig. 2(c), the Q-factor is 1671, 361.6 and 197.1 for Cs = 0.1 pF, 0.5 pF and 1 pF, which presents the resonance sharper with decreasing the series capacitor Cs. In Fig. 2(e), the Q-factor is 4976, 203.4 and 51.85 for Lc = 0.01 nH, 0.05 nH and 0.1 nH, which presents the resonance sharper with decreasing the complementary inductor Lc. In Fig. 2(f), the Q-factor is 831, 712 and 225 for Ls = 20 nH, 15 nH and 5 nH, which presents the resonance sharper with increasing the complementary inductor Ls.

We build the series and parallel resonant circuits parallel in the main-energy thread as shown in Fig. 3(a) and analyze the stable-input impedance:

Figure 3
figure 3

(a) The schematic of the series and parallel resonant circuits are parallel in the main-energy thread. (b) The transmittance of the circuit system in Fig. 3(a) with different parallel-resonant frequencies ωp0 = 0.503 GHz, 50.329 GHz. (c) The transmittance of the circuit system in Fig. 3(a) with different parallel-resonant frequencies ωp0 = 1.9 GHz, 11.18 GHz.

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where the poles expressed as and . Here we maintain the series-circuit elements Ls = 1.0132 nH, Cs =1 pF and thus the series-resonant frequency ωs0 = 5 GHz. Abandoning the physical meaningless solutions, when the series and parallel resonant frequencies satisfying ωp0 ≪ ωs0, the pole ωB of Eq. (5) satisfies ωB ≈ 0 and the other pole ωA is little higher than the zero ωs0. Therefore, the closing of pole and zero can construct transparent-asymmetric and high-Q-factor resonance. Based on the above analysis, we can set the parallel elements Lp = 1.1032 nH, Cp =0.1 pFand thus the parallel-resonant frequency ωp0 = 0.503 GHz. which leads to the transparent resonant frequency ωB = 5.246 GHz closing to the zero ωs0, shown as the dashed line in Fig. 3(b). When the series and parallel resonant frequencies satisfying ωp0 ≫ ωs0, the pole is far from ωs0 and the other pole ωB ≈ ωs0 is little lower than the zero ωs0 in Eq. (5) which constructs the transparent-asymmetric and high-Q-factor resonance. We set the parallel elements Lp = 0.1 nH, Cp = 0.1 pF and thus the parallel-resonant frequency ωp0 = 50.329 GHz, which leads to the transparent resonant frequency ωB = 4.768 GHz closing to the zero ωs0, shown as the solid line in Fig. 3(b). When we set the parallel elements Lp = 1.0132 nH, Cp = 7 pF and parallel-resonant frequency ωp0 = 1.9 GHz. Thus, from the solution of Eq. (5), the pole ωB = 4.753 GHz is located lower than the zero ωs0 which forms the Lorentz-like resonance and the other pole ωA = 6.392 GHz is little higher than the zero ωs0 which forms the transparent-asymmetric and high-Q-factor resonance, shown as the dashed line in Fig. 3(c). When the poles ωs0 − ωB = ωA − ωs0 distribute even around the zero ωs0, the two resonant frequencies locate asymmetric and a sharp reflect-resonance is formed at the zero ωs0 which is likely a converse reversely EIT formation. Here we set the parallel elements Lp = 0.2993 nHand Cp = 5 pF, thus, the the parallel resonant frequency ωp0 = 4.1144 GHz. The transparent resonant frequencies ωA = 5.895 GHz, ωB = 3.49 GHz, which forms two mirror symmetrical resonance, as shown the solid line in Fig. 3(c) and the reflect resonance ωs0 = 5 GHz is also like a reversely EIT phenomenon.

The asymmetric resonance with damping

Here we add the resistors Rs, Rc as damping in the series resonant circuit consisting of the series inductor Ls = 1.0132 nH, the series capacitor Cs = 1 pF and the complementary capacitor Cs = 20 pF, as shown in Fig. 4(a). We set the resistor Rs = 0.1 ohm and Rc = 0.1 ohm respectively, which leads to the amplitude of the transmittance is lower than no-damping, but the shape of the asymmetric resonance remains unchanged, as shown in Fig. 4(b).

Figure 4
figure 4

The schematic of the complementary capacitor Cc = 20 pF parallel in the series LC circuit consisting of the series inductor Ls = 1.0132 nH and the series capacitor Cs = 1 pF with damping which is represented by the resistors Rs and Rc. (b) The transmittance of the circuit system in Fig. 4(a) with different resistors Rs = 0.1 ohm and Rc = 0.1 ohm.

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Conclusion

In conclusion, we show that the Fano resonance can be interpreted as an analogy with the stable-input impedance mechanism by taking passive circuit system as an example. Based on the circuit theory, only three passive components (such as two inductors and one capacitor) can mimic arbitrary Q-factor asymmetric resonance flexiblely by adjusting the pole-zero of the impedance. Furthermore, four passive components can imitate the various resonance (such as Lorentz-like and reversely EIT formations). Besides, our work provides an intuitive understanding of the Fano resonance using briefly electric formation and processes to achieve high-precision by compressed-composition components.

Additional Information

How to cite this article: Lv, B. et al. Analysis and modeling of Fano resonances using equivalent circuit elements. Sci. Rep. 6, 31884; doi: 10.1038/srep31884 (2016).